In the field of radio frequency (RF) communication receivers, the main task of the receiver front-end circuit is to process a signal that is received by an antenna coupled to the receiver front-end circuit in such a manner that it can be more easily processed by subsequent receiver circuits, for example, demodulation circuitry. Typically, such front-end circuits comprise low noise amplifier (LNA) circuitry for amplifying the received RF signal, and mixer circuit arranged to perform frequency translation of the amplified radio frequency signal to a lower intermediate or baseband frequency. The intermediate/baseband frequency signal may then be filtered to remove interfering signals etc.
Since the frequency of the intermediate or baseband signal output by the mixer circuit is typically much lower than the carrier frequency (fRF) for the received RF signal, all stages within the receive chain, subsequent to the mixer circuit, operate at low or baseband frequencies. Furthermore, due to the amplification provided by the LNA circuitry in front of the mixer circuit, and by the mixer circuit itself (if active mixers are used), the signal levels following the mixer circuit are also larger than the signal level of the received RF signal. Accordingly, these low frequency/high signal level characteristics allow the use of a large variety of circuit techniques for the implementation of the stages within the receive chain following the front-end circuitry.
However, due to the high operating frequencies and the low signal levels of the received RF signal, only a very limited number of circuit techniques may be used to successfully implement the front-end circuitry that comprises the LNA circuitry and the mixer circuit. The amplification provided by the LNA increases the signal level at the input of the mixer circuit/device and, therefore, alleviates its noise requirements. However, fully integrated LNAs are known to exhibit poor selectivity. As a consequence, the LNA not only amplifies the wanted signal component, but also amplifies unwanted signal components with frequencies close to the frequency of the wanted signal component. Accordingly, the higher the LNA gain, the more challenging become the linearity requirements for the mixer circuit. A key mixer circuit linearity metric is the IP3 (third order intercept point), since the mixer is often the bottleneck within, say, a receiver front end in terms of IP3, as well as IP2 (second order intercept point).
Referring now to FIG. 1, there is illustrated an example of a known mixer circuit topology in the form of a Gilbert quadrature mixer 100 comprising a pair of Gilbert cell mixers 110, 120. The input stage for each Gilbert cell mixer comprises a respective transconductance stage 115, 125, which transforms the input signal coming from, for example, a preceding LNA or filter into a signal current. This signal current is then chopped by a group of four transistors 130, which down-converts the frequency of the signal current to the desired intermediate or baseband frequency. The output of the mixer 100 is in the form of a current and is typically converted to a voltage by a resistor/capacitor RC load (not shown), which also functions as a low-pass filter.
Two significant limitations of this known mixer design are firstly that the linearity of the mixer is primarily limited by the input transconductance stages 115, 125, and secondly that flicker noise generated by the transistors 130 appears at the output. For the first of these limitations, good mixer designs with a current consumption commensurate to portable devices, such as mobile telephone handsets, etc., require an input-referred IP3 value smaller than circa 0 dBV. Such a stringent requirement is not achievable with the use of such an input transconductance stage. As for the second of the above identified limitations, such a limitation is usually not significant for bipolar transistor implementations, since flicker noise is significantly less for bipolar transistors than it is for, say, MOSFETs (metal oxide semiconductor field effect transistors). However, the presence of flicker noise at the output is a significant problem for CMOS implementations.
Whilst bipolar transistors may be more suitable for implementing mixer circuits in terms of their flicker noise, the fabrication of high performance bipolar transistors requires expensive processing steps during their fabrication. Consequently, the fabrication of such high performance bipolar transistors is prohibitively expensive for cost sensitive implementations, such as within front end circuits of RF communication receivers. Less expensive CMOS processes may be used to produce lower performance bipolar transistors. However, such lower performance bipolar transistors are not capable of operating at the Gigahertz frequencies required by modern RF communication receivers.
A known method for overcoming the flicker noise problem of CMOS implementations of the Gilbert cell mixer is by way of suppressing the DC (direct current) current flowing in the switching transistors, since the flicker noise of CMOS transistors, such as metal oxide semiconductor field effect transistors (MOSFETs), is proportional to the drain bias current flowing in the transistor. FIG. 2 illustrates a known example of a passive Gilbert type mixer 200 whereby coupling capacitors 260 are connected in series with the switching transistors 230 the output ports of the mixer are terminated by transimpedance amplifiers 240, 250, which also function as low-pass filters. In this manner, the coupling capacitors 260 ensure a zero DC bias current flowing in the switching transistors 230, thereby significantly reducing the flicker noise present at the output ports of the mixer. The transimpedance amplifiers are used to hold the source/drain potential of the switching transistors at a known potential. However, since such a mixer design still comprises input transconductance stages 210, 220, the problem of odd-order distortion, and thus poor linearity, is still present.
FIG. 3 illustrates a further example of a known mixer circuit 300 implementing an alternative method for overcoming the flicker-noise problem of the classic Gilbert cell mixer design illustrated in FIG. 1. The mixer circuit 300 of FIG. 3 comprises an active 2LO-LO mixer whereby switches 340 are directly connected to the transconductance stages 310, and are toggled at twice the desired local oscillator frequency (LO) used for output transistors 330. In this manner, the flicker noise of the output transistors 330 does not appear at the output of the mixer circuit. Conversely, the flicker noise of the switches 340 does appear at the output, but as common mode noise, which may be suppressed. However, once again the mixer circuit 300 still comprises input transconductance stage 310, and thus the problem of odd-order distortion, and thus poor linearity, is still present.
FIG. 4 illustrates an example of a further known mixer circuit 400. For each of the known mixer circuits illustrated in FIGS. 1 to 3, the odd-order distortion, and in particular the third order intercept point (IP3) is primarily limited by the input transconductance stages. For the mixer circuit 400 of FIG. 4, there is no input transconductance stage. Instead, capacitors 410 are used to connect the switching transistors directly to an inductor-capacitor (LC) tank 420 used as a load for, say, a preceding LNA. Importantly, the capacitors 410 are constituent parts of the tank 420, and not merely coupling capacitors. In this manner, their impedance and magnitude are relatively significant and will provide some isolation between the respective quadrature I-channel and the Q-channel. The isolation is required in order to keep the noise contribution of transimpedance amplifiers 440 low. In addition, the parallel resonance boosts the signal current coming from the LNA by a factor equal to the quality factor of the LC tank 420. Since the input stage of the mixer circuit 400 comprises only passive components, the linearity of this mixer circuit 400 is superior, as compared to those of FIGS. 1 to 3, since the linearity is only limited by the switching transistors 430 and by the transimpedance amplifiers 440.
A problem with the mixer circuit 400 of FIG. 4 is that the LC tank 420 requires the inclusion of an inductor. Analogue circuits comprising components such as inductors do not scale with improvements in semiconductor manufacturing processes in the same manner as digital circuits. Thus, the presence of analogue components, such as inductors, is a considerable burden on the ability to scale a circuit in order to reduce the size, footprint, etc. of an integrated circuit device.